1. Field of the Invention
The present invention relates to transmission lines for transmitting analog radio-frequency signals of microwave band, millimeter-wave band or the like, or digital signals. More specifically, the invention relates to a transmission line pair including a first transmission line and a second transmission line placed so as to allow itself to be coupled with the first transmission line, and also relates to a radio-frequency circuit including such a transmission line pair.
2. Description of the Related Art
FIG. 17A shows a schematic cross-sectional structure of a microstrip line which has been used as a transmission line in such a conventional radio-frequency circuit as shown above. As shown in FIG. 17A, a signal conductor 103 is formed on a top face of a board 101 made of a dielectric or semiconductor, and a grounding conductor layer 105 is formed on a rear face of the board 101. Upon input of radio-frequency power to this microstrip line, an electric field arises along a direction from the signal conductor 103 to the grounding conductor layer 105, and a magnetic field arises along such a direction as to surround the signal conductor 103 perpendicular to lines of electric force. As a result, the electromagnetic field propagates the radio-frequency power in a lengthwise direction perpendicular to the widthwise direction of the signal conductor 103. In addition, in the microstrip line, the signal conductor 103 or the grounding conductor layer 105 do not necessarily need to be formed on the top face or the rear face of the board 101, but the signal conductor or the grounding conductor layer 105 may be formed within the inner-layer conductor surface of the circuit board on condition that the board 101 is provided as a multilayer circuit board.
The above description has been made on a transmission line for use of transmission of single-end signals. However, as shown in a sectional view of FIG. 17B, two microstrip line structures may be provided in parallel so as to be used as differential signal transmission line with signals of opposite phases transmitted through the lines, respectively. In this case, since paired signal conductors 103a, 103b have signals of opposite phases flow therethrough, the grounding conductor layer 105 may be omitted.
In a conventional analog circuit or high-speed-digital circuit, a cross-sectional structure of which is shown in FIG. 18A and a top view of which is shown in FIG. 18B, two or more transmission lines 102a, 102b are often placed in adjacency and parallel to each other with a high density in their adjoining distance, giving rise to a crosstalk phenomenon between the adjoining transmission lines with the issue of isolation deterioration involved, in many cases. As shown in non-patent document 1, the origin of the crosstalk phenomenon can be attributed to both mutual inductance and mutual capacitance.
Now the principle of occurrence of a crosstalk signal is explained with reference to a perspective view FIG. 19 (a perspective view corresponding to the structure of FIGS. 18A and 18B) of a transmission line pair of two lines placed in parallel and in adjacency to each other with the dielectric substrate 101 assumed as a circuit board. Two transmission lines 102a, 102b are so constructed that the grounding conductor layer 105 formed on the rear face of the dielectric substrate 101 is used as their grounding conductor portions while two signal conductors placed in adjacency and parallel to each other on a top face of the dielectric substrate 101 are used as their signal conductor portions. Assuming that both ends of these transmission lines 102a, 102b are terminated by unshown resistors, respectively, radio-frequency circuit characteristics of the two transmission lines 102a, 102b can be understood by substituting current-flowing closed current loops 293a, 293b for the two transmission lines 102a, 102b, respectively.
Also, as shown in FIG. 19, each of current loops 293a, 293b is made up of a signal conductor which makes a current flow on the top face of the dielectric substrate 101, a grounding conductor 105 on the rear face on which a return current flows, and a resistive element (not shown) which connects the two conductors to each other in a direction vertical to the dielectric substrate 101. It is noted here that the resistive element introduced in such a circuit (i.e., in a current loop) may be not a physical element but a virtual one in which its resistance components are distributed along the signal conductors, where the resistive element may be regarded as one having the same value of characteristic impedance as that of the transmission lines.
Next, the crosstalk phenomenon that would arise upon a flow of a radio-frequency signal in each current loop 293a is concretely explained with reference to FIG. 19. First, as a radio-frequency current 853 flows in the current loop 293a along a direction indicated by an arrow in the figure upon transmission of a radio-frequency signal, a radio-frequency magnetic field 855 is generated so as to intersect the current loop 293a. Since the two transmission lines 102a, 102b are placed in proximity to each other, the radio-frequency magnetic field 855 intersects even the current loop 293b of the transmission line 102b, so that an induced current 857 flows in the current loop 293b. This is the principle of development of a crosstalk signal due to mutual inductance.
Based on this principle, the induced current 857 generated in the current loop 293b flows toward a near-end side terminal (i.e., a terminal in an end portion on the front side in the figure) in a direction opposite to the direction of the radio-frequency current 853 in the current loop 293a. Since intensity of the radio-frequency magnetic field 855 depends on the loop area of the current loop 293a and since intensity of the induced current 857 depends on the intensity of the radio-frequency magnetic field 855 intersecting the current loop 293b, the crosstalk signal intensity increases more and more as a coupled line length Lcp of the transmission line pair composed of the two transmission lines 102a, 102b increases.
Further, another crosstalk signal is induced to the transmission line 102b due to the mutual capacitance occurring to between the two signal conductors as well. The crosstalk signal generated by the mutual capacitance has no directivity, and occurs to both far-end and near-end sides each at an equal intensity. The crosstalk phenomenon occurring on the far-end side can be construed as a sum of the above two phenomena. Now, current elements generated in the transmission line pair in accompaniment to the crosstalk phenomenon during transmission of high-speed signals are shown in a schematic explanatory view of FIG. 20. As shown in FIG. 20, when a voltage Vin is applied to a terminal 106a on the left side of the transmission line 102a as in the figure, a radio-frequency current element Io flows through the transmission line 102a due to a radio-frequency component contained at a pulse leading edge. A difference between a current Ic generated due to a mutual capacitance by this radio-frequency current element Io and a current Ii generated due to the mutual inductance flows as a crosstalk current into a far-end side crosstalk terminal 106d of the adjacently placed transmission line 102b. On the other hand, a crosstalk current corresponding to the sum of currents Ic and Ii flows into a near-end side crosstalk terminal 106c. Under such a condition that paired transmission lines are placed in proximity to each other at a high density, the current Ii is generally higher in intensity than the current Ic, and therefore a crosstalk voltage Vf of the negative sign, which is inverse to the sign of the voltage Vin applied to the terminal 106a is observed at the far-end side crosstalk terminal 106d. In addition, a voltage Vout is observed at a terminal 106b of the transmission line 102a. 
Here is explained a typical example of crosstalk characteristics in conventional transmission lines. For example, as shown in FIGS. 18A and 18B, on a top face of a dielectric substrate 101 of resin material having a dielectric constant of 3.8, a thickness H of 250 μm and having a grounding conductor layer 105 provided over its entire rear face, is fabricated a radio-frequency circuit having a structure that two signal conductors, i.e. transmission lines 102a and 102b, with a wiring width W of 100 μm are placed in parallel with a wire-to-wire gap G set to 650 μm, where one radio-frequency circuit defined here and having a coupled line length of 50 mm is assumed as Prior Art Example 1 and another of 500 mm as Prior Art Example 2 (it is noted that Prior Art Example 2 will be mentioned later). A wiring distance D, which is a placement distance of the two transmission lines 102a, 102b, is G+(W/2)×2=750 μm. It is noted that those signal conductors are provided each by a copper wire having an electrical conductivity of 3×108 S/m and a thickness of 20 μm.
With respect to such a radio-frequency circuit of Prior Art Example 1, forward transit characteristics by four terminal measurement (terminal 106a to terminal 106b) as well as far-end directed isolation characteristics (terminal 106a to terminal 106d) are explained below with reference to a graph-form view showing the frequency dependence of the isolation characteristics about the radio-frequency circuit of Prior Art Example 1 shown in FIG. 21. It is noted that in the graph of FIG. 21, the horizontal axis represents frequency (GHz) and the vertical axis represents a transit intensity characteristic S21 (dB) and isolation characteristic S41 (dB).
As shown by the isolation characteristic S41 of FIG. 21, the crosstalk intensity monotonously increases with increasing frequency. More specifically, it can be understood that even an isolation of 11 db with the frequency band of 5 GHz or higher, or 7 db with the frequency band of 10 GHz or higher, or as small as 3 db with the frequency band of 20 GHz or higher cannot be ensured. Furthermore, as longer the coupled line length Lcp becomes, or as the placement distance D is decreased, the crosstalk intensity monotonously increases.
Also, as shown by the transit intensity characteristic S21 (indicated by thin line in the figure) of FIG. 21, as the crosstalk signal intensity increases, the transit signal intensity extremely lowers. Specifically, there occurs a decrease of as much as 9.5 db in the signal intensity at 25 GHz. In the radio-frequency circuit of Prior Art Example 1, with transit through a line length of 50 mm, a transit phase of a signal having a frequency of about 1.8 GHz corresponds to 180 degrees. The crosstalk intensity at this frequency is −21.4 db. Although depending on the placement distance D, the crosstalk phenomenon matters in frequency bands in which the coupled line length Lcp corresponds effectively to a wavelength order, i.e. an effective line length of half-wave length or more. For example, decreasing the placement distance D to 200 μm causes the crosstalk intensity to become −15.8 db, and the extending the placement distance D to 1000 μm cause the crosstalk intensity to become 26.7 db. Also, with the placement distance D equal to 200 μm, it becomes impossible to maintain a crosstalk intensity of −10 dB at a frequency of 11.6 GHz at which the coupled line length Lcp corresponds to about 2.5 times the effective wavelength. Also with the placement distance D equal to 750 μm, a crosstalk intensity of −10 db is recorded at a frequency of 25.7 GHz at which the coupled line length Lcp corresponds to about 7 times the effective wavelength. Thus, although depending on the degree of coupling between lines, influences of the crosstalk phenomenon becomes quite considerable under the condition that the coupled line length Lcp corresponds to a double or more of the effective wavelength.
As a conventional technique purposed to suppress such a crosstalk phenomenon, there has been a transmission line structure shown in patent document 1 as an example. The transmission line structure shown in patent document 1 is a structure which is effective for optimizing the electromagnetic field distribution of high frequencies during signal transmission to reduce the crosstalk about a unit line length. That is, since it is the coupling between parallel lines described above that makes the factor of the crosstalk, this is a technique intended to suppress the crosstalk phenomenon by providing a transmission line cross-sectional structure which is so designed as to reduce the degree of coupling between parallel lines. More specifically, as shown in a cross-sectional structure of a transmission line pair of FIG. 22, a second dielectric 145 which is lower in dielectric constant than a first dielectric 144 serving as the substrate is distributed at a partial site of the substrate between two signal conductors 142 and 143 of the transmission line pair. Since the radio-frequency electric field intensity of the signal traveling on the transmission lines is lowered at the distribution site of the second dielectric 145 of low dielectric constant, the degree of coupling between the transmission lines can be lowered, thus making it achievable to suppress the crosstalk phenomenon.                Patent document 1: Japanese Unexamined Patent Publication No. 2002-299917 A        Patent document 2: Japanese Unexamined Patent Publication No. 2003-258394 A        Non-patent document 1: An introduction to signal integrity (CQ Publishing Co., Ltd., 2002) pp. 79        